Narrowband interference suppression for ofdm system

ABSTRACT

A method for suppressing narrowband interference in OFDM receivers is provided including the steps of acquiring a sample of received data, estimating parameters of each of a number of narrowband interferers from the acquired sample of data, forming an excision filter using the estimated parameters and inserting the excision filter into an OFDM receiver.

FIELD OF INVENTION

The invention relates to narrowband interference suppression systems forOFDM communications system and in particular to excision filtering fornarrowband interference suppression system for OFDM communicationsystems.

BACKGROUND

Orthogonal frequency division multiplexing (OFDM) has become thephysical layer of choice for many wireless communications systems. Anattractive feature of current wireless local area network (WLAN) andwireless metropolitan area network (WMAN) standards based on OFDM is thedesigned ability to operate in unlicensed spectrum. However, thesesystems must share spectrum with other unlicensed systems; such ascordless telephones, garage door openers, baby monitors and microwaveovens; which produce narrowband interference in WLAN and WMAN systems.Further, radio non-idealities such as transmitter carrier feedthrough(also known as carrier leakage) also introduce narrowband interferencein the form of single-frequency carrier residues.

Pilot symbol assisted systems are particularly susceptible to narrowbandinterference during receiver detection and synchronisation. Pilot symbolassisted systems are also susceptible to narrowband interference on thedata transport phase of receiver operation. An interference suppressiontechnique has been proposed to improve the performance of pilot symbolassisted detection and synchronisation in the presence of narrowbandinterference, see PCT/NZ2004/000060. However, this technique cannot beapplied during data transport as it introduces inter-symbolinterference.

Previous proposed interference suppression systems for OFDM includeusing pre-coding, spread spectrum OFDM, and post-detection receivertechniques involving equalizers. There are many literature reports onnarrowband interference suppression techniques for spread spectrumsystems, including excision-based methods. However the interferencesuppression requirements for OFDM differ significantly from therequirements for spread spectrum.

Previously proposed methods for narrowband interference suppressioninclude using pre-coding and spread spectrum techniques. These methodsrequire modifications to the transmitted OFDM signal which are notsupported by current OFDM standards. Frequency domain techniques alsohave been proposed, but these either do not take account of spectralleakage from the interference, or require substantial processing toestimate and remove the interference from all OFDM frequency bins.Additionally, another proposed method requires co-operation of thetransmitter in not sending data on a number of sub-carriers in thevicinity of (that is, close to in frequency) each narrowband interferer.Again, this ability to modify the transmitted signal is not supported bythe current standards.

SUMMARY OF INVENTION

It is the object of the present invention to provide an improved systemand method for interference suppression in OFDM communications systemsor to at least provide the public with a useful choice.

In broad terms in one aspect the invention comprises a method forsuppressing narrowband interference in OFDM receivers including thesteps of acquiring a sample of received data, estimating parameters ofeach of a number of narrowband interferers from the acquired sample ofdata, forming an excision filter using the estimated parameters andinserting the excision filter into an OFDM receiver.

Preferably the filter is inserted into the OFDM receiver prior to adiscrete Fourier transform.

Preferably the estimated parameters of the narrowband interferersinclude demodulated carrier frequency, magnitude and phase.

Preferably the step of estimating the number of narrowband interferersincludes the steps of performing a forward DFT on the samples, andperforming a periodogram search on the output of the DFT to identifypeaks in the periodogram where the number of peaks in the periodogramcorresponds to the number of interferers.

Preferably the step of estimating parameters of the narrowbandinterferers includes estimating the location of an interferer as thefrequency of a peak on the corresponding periodogram, estimating themagnitude of the interferer as the amplitude of the correspondingperiodogram peak, and estimating the phase of the interferer as thephase of the corresponding periodogram peak.

Preferably the narrowband interferer parameter estimates of eachnarrowband interferer are used to initialise a digital phase lockedloop.

Preferably the method for suppression narrowband interference includesthe step of receiving an indication of a start of packet when a datapacket is received by the OFDM receiver.

Preferably the phase locked loops are updated with each incoming sampleuntil either a counter expires or an OFDM packet is detected. The phaselocked loops are used to estimate the carrier frequency of thenarrowband interferers. Preferably the phase locked loops are digitalphase locked loops. Preferably one phase locked loop is used for eachinterferer.

Preferably the current narrowband interferer carrier frequency estimatesfrom the phase locked loops that have achieved “lock” are used toinitialise an excision filter when an OFDM packet is detected.

The excision filter may have impulse response duration less than theOFDM guard interval.

In broad terms in a further aspect the invention comprises an OFDMreceiver including a front end arranged to receive data, a data samplerarranged to provide samples of received data, a narrowband interferencedetector that detects narrowband interferers in the sample of receiveddata and estimates parameters of each narrowband interferer, and anexcision filter that uses the estimated parameters of each narrowbandinterferer to reduce noise from the narrowband interferers.

Preferably the excision filter is inserted into the OFDM receiver priorto a Fourier transform operator.

Preferably the narrowband interference detector estimates thedemodulated carrier frequency, magnitude and phase of the narrowbandinterferers.

Preferably the narrowband interference detector includes a Fouriertransform operator arranged to perform a Fourier transform on thesamples and perform a periodogram search on the output of the Fouriertransform operator to identify peaks in the periodogram and at least onephase lock loop arranged to lock onto a peak identified by theperiodogram search.

Preferably the narrowband interference detector is further arranged toestimate the frequency of an interferer as the location of a peak on thecorresponding periodogram, estimate the magnitude of the interferer asthe amplitude of the corresponding periodogram peak, and estimate thephase of the interferer as the phase of the corresponding periodogrampeak.

Preferably the narrowband interference detector includes a timer and afilter design module.

Preferably the OFDM receiver is further arranged to provide an estimateof the start of an OFDM data packet to the narrowband interferencedetector.

Preferably the narrowband interference detector is arranged to innovatethe phase lock loop(s) until either the timer times out or an OFDMpacket is received.

Preferably the phase locked loops are arranged to estimate the carrierfrequency of the narrowband interferers.

Preferably one phase locked loop is used for each interferer.

Preferably the current narrowband interferer carrier frequency estimatesfrom the phase locked loops that have achieved “lock” are used by thefilter estimator to initialise an excision filter when an OFDM packet isdetected.

The excision filter may have impulse response duration less than theOFDM guard interval.

BRIEF DESCRIPTION OF DRAWINGS

The invention will be further described by way of example only andwithout intending to be limiting with reference to the followingdrawings, wherein:

FIG. 1A shows the bit error rate performance of BPSK modulated OFDM witha single interferer and signal to interference ratio (SIR) of −10 dB;

FIG. 1B shows the bit error rate performance of BPSK modulated OFDM witha single interferer and SIR of 10 dB;

FIG. 2A shows an interference suppression detector of the invention;

FIG. 2B is a flowchart showing one technique for interferencesuppression of the invention;

FIG. 3 is a block diagram of an OFDM receiver including the interferencesuppression system of the invention;

FIG. 4A shows a simulation of interferer carrier frequency estimationwhere the INR is 7.5 dB;

FIG. 4B shows a simulation of interferer carrier frequency estimationwhere the INR is 6.6 dB;

FIG. 4C shows phase lock loop indication for the interferers of FIGS. 4Aand 4B;

FIG. 5A shows a prototype excision filter frequency response;

FIG. 5B shows an example of a two notch excision filter;

FIG. 6A shows an example of a received signal with the narrowbandinterference suppression system in place;

FIG. 6B shows the smoothed spectra of a first OFDM data block afternarrowband interference is suppressed;

FIG. 7A shows a received signal constellation including signal,narrowband interference and noise;

FIG. 7B shows the received signal constellation after filtering toremove narrowband interference;

FIG. 7C shows the received signal constellation when no narrowbandinterference is present;

FIG. 8A is a simulation of bit error rates for BPSK modulated OFDM withtwo narrowband interferers and SIR of −10 dB;

FIG. 8B is a simulation of bit error rates for BPSK modulated OFDM withtwo narrowband interferers and SIR of 0 dB;

FIG. 8C is a simulation of bit error rates for BPSK modulated OFDM withtwo narrowband interferers and SIR of 10 dB;

FIG. 9A is a simulation of bit error rates for BPSK modulated OFDM withtwo narrowband interferers and SNR of 6 dB;

FIG. 9B is a simulation of bit error rates for BPSK modulated OFDM withtwo narrowband interferers and SNR of 12 dB;

FIG. 9C is a simulation of bit error rates for BPSK modulated OFDM withtwo narrowband interferers and SNR of 18 dB;

FIG. 10A is a simulation of bit error rates for 64-QAM modulated OFDMwith two narrowband interferers and SIR of 0 dB; and

FIG. 10B is a simulation of bit error rates for BPSK modulated OFDM withtwo narrowband interferers and SIR of −15 dB.

DETAILED DESCRIPTION

A common model for a received, baseband (low pass equivalent) OFDMsymbol, sampled with period T, isr _(n) =c(τ;nT){circle around (×)}s(nT−τ _(s))e ^(−j[2πv(nT−r) ^(s)^()+θ])+η(nT)   (1)where c(τ;nT) is a doubly dispersive, low pass equivalent, fadingchannel which introduces time-dispersion in dimension τ, s(t) is thetransmitted signal, n is the sample index, τ_(s), v and θ are,respectively, the time-, frequency- and phase-offsets betweentransmitter and receiver introduced by a combination of systemnon-idealities and channel linear distortions, and η is complex additivewhite Gaussian noise (AWGN) having variance σ_(w) ² and {circle around(×)} is the convolution operator. Addition to N narrowband interferes tothe received signal produces $\begin{matrix}{r_{n} = {{{{c\left( {\tau;{n\quad T}} \right)} \otimes {s\left( {{n\quad T} - \tau_{s}} \right)}}{\mathbb{e}}^{- {j{\lbrack{{2\pi\quad{v{({{nT} - \tau_{s}})}}} + \theta}\rbrack}}}} + {{c\left( {\tau;{n\quad T}} \right)} \otimes {\sum\limits_{i = 1}^{N}{b_{i}{\mathbb{e}}^{- {j{\lbrack{{2{\pi\xi}_{i}{nT}} + \phi_{i}}\rbrack}}}}}} + {\eta\left( {n\quad T} \right)}}} & (2)\end{matrix}$where b_(i), ξ_(i) and φ_(i) are, respectively, the amplitude, frequencyand phase of the ith of N demodulated narrowband interferer.

This simple interference model is realistic for narrowband FM (egcordless telephones, baby monitors), for low rate digital modulators (eggarage door openers), and for carrier feedthrough. For other interferersof greater bandwidth (eg microwave ovens and Bluetooth devices), morecomplex models than equation (2) may be required. Carrier feedthrough inthe transmitter produces an in-band interferer at a frequency equal tothe frequency difference between transmitter and receiver localoscillators which, depending on the amount of Doppler shift, will beequal or close to the frequency offset, v. Typically, the maximumcarrier frequency offset is much less than the OFDM sub-carrier spacingand the pilot symbol is designed specifically to be able to resolve thisfrequency without ambiguity. Any DC offset will occur at narrowbandinterference frequency, ξ=0 and interference from other users oflicense-free spectrum may occur either singly (e.g. garage door openers,baby monitors, microwave ovens) or in pairs (e.g. cordless telephones)at any in-band frequency. It is noted that, at the time of writing,anecdotal evidence suggests a much higher likelihood of interference inthe 2.4 GHz ISM band than in the 5 GHz band. This suggests that theprincipal immediate application of the invention maybe to IEEE 802.11g-compliant WLANs.

Each narrowband interferer will experience only frequency flat multipathfading, so the effect of the fading channel on the narrowbandinterferers may be incorporated by modifying the random amplitude andphase of each. Further, assuming that the cyclic prefix of each OFDMblock has been chosen to be sufficiently long so as to preventinter-symbol interference, then the effect of the fading channel on theOFDM signal may be treated as being purely multiplicative on a per blockbasis. Since the focus of this invention on bit errors produced bynarrowband interference, the multiplicative effect of the fading channelon each OFDM signal sub-carrier is neglected and frequency flat fadingalso is assumed to apply to the OFDM signal. Moreover, as the primaryconcern here is with packet-based systems where the maximum packetduration is very much shorter than the typical operating channelcoherence time, the effect of time variation in the channel also mayneglected. Where the validity of each of these assumptions becomesimportant, this is commented on at appropriate points below. It isassumed also that packet detection and synchronisation have beenachieved. Thus, ideal, uncoded systems which are perfectly synchronisedare considered: practical system performance is dependent on thespecific detection, synchronisation and coding/decoding algorithmsemployed—these are not within the scope of this paper. Incorporation ofthese assumptions simplifies the model to $\begin{matrix}{{r_{n} = {{a\quad s_{n}} + {\sum\limits_{i = 1}^{N}{b_{i}{\mathbb{e}}^{- {j{\lbrack{{2{\pi\xi}_{i}{nT}} + \phi_{i}}\rbrack}}}}} + \eta_{n}}},} & (3)\end{matrix}$where a is the static Gaussian channel attenuation.

At the transmitter, an OFDM symbol is produced such that $\begin{matrix}{{s_{n} = {w_{n}{\sum\limits_{k = 0}^{L - 1}{d_{k}{\mathbb{e}}^{{j2\pi}\quad n\frac{k}{L}}}}}},} & (4)\end{matrix}$is the nth of L samples in the OFDM symbol, where d_(k) is the kth datasymbol from some modulation constellation (for example, m-PSK or m-QAM),and w_(n) is a windowing function which is often simply rectangular(w_(n)=1, ∀n). Typically, a K member subset of {d_(k)} is set to zero asspectral blanking, thus L time domain samples represent L-K frequencydomain symbols.

At the receiver, the L-point inverse discrete Fourier transform (DFT) ofthe narrowband interferer has the kth sample $\begin{matrix}{\begin{matrix}{I_{k} = {\sum\limits_{n = 0}^{L - 1}{b\quad{\mathbb{e}}^{- {j{\lbrack{{2{\pi\xi}\quad{nT}} + \phi}\rbrack}}}{\mathbb{e}}^{{- {j2\pi}}\quad k\frac{n}{L}}}}} \\{{= {b\quad{\Psi_{k}\left( {\xi,\phi} \right)}}},}\end{matrix}{where}} & (5) \\{{\Psi_{k}\left( {\xi,\phi} \right)} = {{\mathbb{e}}^{- {j{\lbrack{{{\pi{({L - 1})}}{({\frac{k}{L} - {\xi\quad T}})}} + \phi}\rbrack}}}\frac{\sin\quad\pi\quad{L\left( {\frac{k}{L} - {\xi\quad T}} \right)}}{\sin\quad\pi\quad\left( {\frac{k}{L} - {\xi\quad T}} \right)}}} & (6)\end{matrix}$is the sampled circular sinc function centred at the interfererfrequency. Where the interferer frequency does not coincide exactly witha DFT frequency sample (that is, where ξT≠k/L,k∈{0. . . L−1}) then|I_(k)|>0,∀k, which is known as spectral leakage. Thus, the amount ofinterference experienced by each data symbol depends on the particularvalue of the interferer frequency.

For single interferers having particular values of received amplitude b,normalised carrier frequency ξT and phase φ of the (receiver) mean BERin the k^(th) BPSK modulated OFDM data bin is given by the conditionaldistribution $\begin{matrix}{{p_{e}\left( {\gamma_{b},{\gamma_{i}❘k},\xi,\phi} \right)} = \left\lbrack \quad{{\frac{1}{4}{erf}\quad{c\left( {\sqrt{\gamma_{b}} + {\sqrt{\gamma_{i}}{Re}\left\{ {\Psi_{k}\left( {\xi,\phi} \right)} \right\}}} \right)}} + {\frac{1}{4}{erf}\quad{c\left( {\sqrt{\gamma_{b}} - {\sqrt{\gamma_{i}}{Re}\left\{ {\Psi_{k}\left( {\xi,\phi} \right)} \right\}}} \right)}}} \right\rbrack} & (7)\end{matrix}$

where γ_(b)=Ad²/σ_(W) ² is the mean SNR per bit for (mean) BPSK bitenergy d², γ_(i)=b²/σ_(W) ² is the mean INR per sample, and σ_(W) ² isthe noise power per bit. Note that A and σ_(W) ² are frequency domainduals of the time domain quantities a and σ_(w) ² expressed in equations(1) and (3). This expression is derived from geometric considerationsnoting that the summation of two complementary error functions averagesthe BER for two possible values of the BPSK data bit.

The mean BER across each BPSK modulated OFDM symbol for a singleinterferer having particular values of ξT and φ is obtained by averagingacross all data bins, thus $\begin{matrix}{{{p_{e}\left( {\gamma_{b},{\gamma_{i}❘\xi},\phi} \right)} = {\frac{1}{L - K}{\sum\limits_{k = 0}^{L - K - 1}{p_{e}\left( {\gamma_{b},{\gamma_{i}❘k},\xi,\phi} \right)}}}},} & (8)\end{matrix}$noting that only L-K data bins in the OFDM symbol carry data. Note alsothat the effect of frequency-selective fading additionally can beaccounted for in equation (8) by replacing frequency flat A with A_(k),the channel gain per frequency bin. Insertion of particular values ofinterferer frequency, ξT, and interferer phase, φ, into equation (8)demonstrates there is considerable variation in mean BER per OFDM symbolacross the ensemble of ξT and φ. Thus, for a given interferer, theparticular values of interferer phase, ξT, and interferer phase, φ,largely determine the BER for an individual received packet. This isillustrated in FIGS. 1A and 1B, which show analytical bit error ratesfor particular values of ξT and φ, as well as for the ensemble average(both analytical and simulation) and for interference-free (ideal) BPSKmodulated OFDM.

The ensemble average BER for BPSK modulated OFDM in the presence of asingle narrowband interferer can be found by integrating equation (8)over the ensemble of ξT and φ, producing $\begin{matrix}\begin{matrix}{{p_{e}\left( {\gamma_{b},\gamma_{i}} \right)} = {\int_{k}{\int_{\xi}{\int_{\phi}{{p_{e}\left( {\gamma_{b},{\gamma_{i}❘k},\xi,\phi} \right)}{p_{k}(k)}{p_{\xi}(\xi)}{p_{\phi}(\phi)}{\mathbb{d}\phi}{\mathbb{d}\xi}{\mathbb{d}k}}}}}} \\{= {\frac{1}{L - K}{\sum\limits_{k = 0}^{L - K - 1}{\frac{1}{2\pi}{\int_{- \pi}^{\pi}{T{\int_{0}^{\frac{1}{T}}{{p_{e}\left( {\gamma_{b},{\gamma_{i}❘k},\xi,\phi} \right)}{\mathbb{d}\phi}{{\mathbb{d}\xi}.}}}}}}}}}\end{matrix} & (9)\end{matrix}$

Expressions similar to equation (9) have been developed previously forevaluating the BER of direct sequence spread spectrum (DSSS) sufferingnarrowband interference. The principal difference is that the previouslydeveloped expressions average SNR and INR across frequency within theerror function, whereas equation (9) averages SNR and INR acrossfrequency outside the error function. This highlights that BER for DSSSdepends on average narrowband interference power, thus is insensitive toparticular values of interferer carrier frequency and phase, and so anytechniques that reduce average SIR will improve BER performance. Bycontrast, BER for OFDM depends on interference power per DFT bin, thusis sensitive to particular values of interferer carrier frequency andphase, and the DFT itself increases BER by introducing interference tomore data-bearing carriers through spectral leakage. This is illustratedin FIGS. 1A and 1B which compare the ensemble average BERs of OFDM andDSSS for a single narrowband interferer. Note that the SNR andsignal-to-interference ratio (SIR) are both expressed as average persample quantities in FIGS. 1A and 1B.

As can be seen from these Figures the bit error rates are highest forthe interferer where the interferer frequency, ξT, is not equal to theDFT frequency sample, K/L, and the phase of the interferer is π/2. Itshould be noted that the legend in FIG. 1B applies also to FIG. 1A.These figures show close agreement of ensemble averages producedanalytically and by computer simulation. Note also the variation withinthe ensemble, indicated both by analytical BER curves for particularvalues within the ensemble and by the 10^(th) and 90^(th) percentilelimits on the computer simulation BER curves (shown as error bars).

Of particular importance in FIGS. 1A and 1B is the behaviour of theensemble average BER of BPSK modulated OFDM in the presence of a singlenarrowband interferer. FIG. 1A shows that, for an SIR of −10 dB, themean BER asymptotes at about 0.06 at which level even the heaviestcoding/decoding may fail to produce an error-free packet. Note that, forSNRs of greater than 10 dB, the mean BER is independent of SNR. FIG. 1Bshows that, even for a modest SIR of 10 dB, the mean BER asymptotes atabout 0.008, requiring heavy coding/decoding to produce error freepackets.

For multiple narrowband interferers, the mean BPSK modulated BER in thekth OFDM data bin can be obtained by modifying equation (7) to produce$\begin{matrix}{{p_{e}\left( {\gamma_{b},{{\gamma_{1}\quad\ldots\quad\gamma_{N}}❘k},\xi,\phi} \right)} = \left\lbrack {{\frac{1}{4}{erf}\quad{c\left( {\sqrt{\gamma_{b}} + {\sum\limits_{i = 1}^{N}{\sqrt{\gamma_{i}}{Re}\left\{ {\Psi_{k}\left( {\xi_{i},\phi_{i}} \right)} \right\}}}} \right)}} + {\frac{1}{4}{erf}\quad{c\left( {\sqrt{\gamma_{b}} - {\sum\limits_{i = 1}^{N}{\sqrt{\gamma_{i}}{Re}\left\{ {\Psi_{k}\left( {\xi_{i},\phi_{i}} \right)} \right\}}}} \right)}}} \right\rbrack} & (10)\end{matrix}$where γ_(i)=b_(i) ²/σ_(W) ² here is the mean INR per sample for the ithinterferer. Respective average BER expressions for multiple interfererscan be obtained by inserting equation (10) into equation (8) and (9) asrequired.

Excision-based methods of interference cancellation require estimationof the carrier frequency only (but not amplitude or phase) for eachnarrowband interferer. For an excision filter having a frequencyresponse H(f), the post-excision conditional BER for BPSK modulated OFDMis produced by modifying equation (7) to obtain $\begin{matrix}{{{p_{e}\left( {\gamma_{b},{\gamma_{i}❘k},\xi,\phi} \right)} = \left\lbrack {{\frac{1}{4}{erf}\quad{c\left( {\sqrt{\gamma_{b}} + {\frac{H(\xi)}{H\left( \frac{k}{T} \right)}\sqrt{\gamma_{i}}{Re}\left\{ {\Psi_{k}\left( {\xi,\phi} \right)} \right\}}} \right)}} + {\frac{1}{4}{{erfc}\left( {\sqrt{\gamma_{b}} - {\frac{H(\xi)}{H\left( \frac{k}{T} \right)}\sqrt{\gamma_{i}}{Re}\left\{ {\Psi_{k}\left( {\xi,\phi} \right)} \right\}}} \right)}}} \right\rbrack},} & (11)\end{matrix}$which may be inserted into equation (9) to produce the marginaldistribution representing the ensemble BER. Each OFDM sub-channel forwhich the frequency response of the OFDM data is greater than thefrequency response of the narrowband interferer,${{H\left( \frac{k}{T} \right)} > {H(\xi)}},$will experience an improved BER compared to the unfiltered equivalent.In particular when the frequency response of the narrowband filter iszero, H(ξ)=0, equation (11) reduces to the ideal BER for BPSK notingthat, due to finite precision in any practical digital implementationthis condition is seldom met. Note that equation (11) is valid only forexcision filters having impulse response duration less than the OFDMguard interval. Where this is not the case, the excision filter willintroduce some degree of inter-symbol interference.

The main challenges for excision filtering are, firstly, estimation ofthe interferer carrier frequency (such that the notch of the excisionfilter is correctly positioned in frequency) and, secondly, design andimplementation of an effective and efficient notch filter. To accountexplicitly for the effect of carrier frequency estimation error onequation (11), the frequency response of the narrowband interferer,H(ξ), can be replaced with H(ξ|ε_(ξ)), the excision filter frequencyresponse at the true interferer frequency conditioned on the filterhaving been designed to excise the estimated interferer frequency, whereε_(ξ)=ξ−{circumflex over (ξ)} is the particular value of estimationerror and {circumflex over (ξ)} is the estimated frequency of thenarrowband interferer. The marginal distribution of equation (9) mustthen be integrated additionally over the estimation error distributionp₆₈ (ε_(ξ))dε_(ξ) to obtain the ensemble average BER. Note that excisionfiltering does not require the estimation of interferer power, so anytime-variation introduced by the fading channel will not reduce theefficacy of excision filtering. Also, provided that the filter notchwidth is sufficient, excision filtering is insensitive to some timevariation in interferer frequency, due to either modulation of theinterferer or time variation in the channel.

FIG. 2A is a block diagram of the interference suppression technique ofthe invention and FIG. 2B is a flow chart of the interferencesuppression technique of the invention. The narrowband interferencesuppression system can run in parallel with a technique for interferencesuppression during pilot symbol assisted detection and synchronisationsuch as that described in the applicant's patent applicationPCT/NZ2004/000060 which is herein incorporated by reference. Themultiple narrowband interference suppression system of the inventionrelies on estimating interference carrier frequency(s) during thesignal-free period between data packets—this reduces the impact of theparticular values of SIR and interferer carrier frequency on thealgorithm performance. The estimated carrier frequencies are used tospecify the excision filter applied to the received signal afterdetection. The multiple narrowband interference suppression system ofthe invention is particularly suited for interference suppression duringthe data transport phase of the data reception. However this techniquecan also be used during packet detection and synchronisation.

FIG. 2B shows the steps of one embodiment of the excision-basednarrowband interference suppression technique. When detection is startedin step 1 an initial block of L samples of interference plus noise (noOFDM signal) is collected. The number of samples, L, is the block sizeof the available DFT software/firmware. In preferred embodiments L istypically the OFDM block size.

After the block of L is collected in step 2 maximum likelihoodparameters estimates for the desired narrowband interferers areperformed using the forward DFT on the block of L samples. After takinga DFT of the data block an assumption is made of the number ofnarrowband interferers present. Assuming that M interferers areanticipated as being present, then a periodogram search (whichdetermined arg(max([S₀ . . . S_(L-1)])) for DFT output samples (S₀ . . .S_(L-1))) is employed to identify the M largest periodogram peaks fromthe DFT output. To avoid false detection of interferers due to spectralleakage, periodogram bins either side of each identified peak are notconsidered in subsequent (recursive) searches. The assumption of thenumber of interferers present is based on the maximum number ofinterferers that could be present. For example, if interference iscaused by a cordless telephone there could be two interferers presentbut it is less likely that more than two interferers will be present.There is a performance/complexity trade-off in the number of interferersestimated to be present. Estimating a high number of interferers willgive better performance but will also increase the complexity of thesystem compared to estimating a lower number of interferers.

After the periodogram search has identified the amplitude and phase ofeach identified periodogram peak are the estimated interferer magnitudeand phase, respectively as shown in step 3. The frequency correspondingto the peak periodogram location is the estimated interferer demodulatedcarrier frequency.

In step 4 the M parameter estimate vectors are used to initialise Mdigital phase locked loops (PLLs). At this point a time-out counter isset.

In step 5 the equation is asked whether a packet has been detected. Inthe embodiment packet detection is performed by another function (shownas step 12). If no packet has been detected the ‘no’ arrow is followedfrom step 5 to step 10. If a packet has been detected the ‘yes’ arrow isfollowed from step 5 to step 6 and packet reception commences.

In step 10 the question is asked has the time-out counter expired. Thetime-out counter is used to ensure that the interferer estimates arecurrent. For example, if a new interferer appears after the block of Lsamples is taken in step 1 it will not be compensated for unlessperiodically the process repeats by using the latest block of L samplesto estimate the narrowband interferer parameters. In this way thetechnique should track the M strongest narrowband interferers. If thetime-out counter has not expired the ‘no’ arrow is followed to step 11.If the time-out counter has expired the ‘yes’ arrow is followed to step2 and the latest block of L samples is used to estimate the narrowbandinterferers. This ensures that new interferers are identified as theyappear. In this way, the invention should track the M strongestnarrowband interferers.

When the ‘no’ arrow is followed to step 11 the M PLLs are innovated andthe time-out counter is incremented. The arrow is then followed to step5. In this way the M PLLs are innovated with each incoming sample untileither an OFDM packet is detected by a parallel process or the time-outcounter expires. Each PLL locks onto an interferer to provide thecarrier frequency of the interferer. If a PLL lock is not achieved thenit can be assumed that no interferer is present for that PLL. The numberof locks achieved indicates the number of interferers present. Forexample if two PLLs are initialised and innovated and only one achievesa lock it can be assumed that only one interferer is present. When thisoccurs only one excision filter is produced as the second interferer (ifpresent) is deemed to have negligible impact on the BER.

When a packet is detected the ‘yes’ arrow is followed from step 5 tostep 6. If a packet has been detected, then the current M interfererparameter estimates are used to initialise the excision filter in step7. Importantly, each PLL provides a lock indication (for exampleI_(lock)=Im(ε_(PLL))/Re(ε_(PLL)), whereε_(PLL)=r_(n)e^(j[2π{circumflex over (ξ)}nT+{circumflex over (φ)}]) isthe PLL error for the received signal, noise and narrowbandinterference, r_(n), defined by equation (3) with the transmitted signals_(n)=0, approaches zero when phase lock has been achieved) which allowsa genuine interferer to be distinguished from thermal noise. Onlyinterferers for which PLL lock is achieved are excised from the OFDMpacket.

During data reception, interferers are removed using one or moreexcision (notch) filter(s) centred at the estimated interfererdemodulated carrier frequency shown in step 8. This filter is insertedat the input to the OFDM digital receiver prior to the forward DFT toprevent spectral leakage.

FIG. 2A is a block diagram of the devices used in the interferencesuppression technique of the invention. The interference suppressionapparatus of the invention is shown in blocks 33 and 34 and ispositioned between front end 31 and OFDM receiver 35. The make-up of thefront end 31 and the OFDM receiver 35 are described in more detail inFIG. 3. The narrowband interference detector includes an N bin FFT 36, atimer/counter 37, switches 38 and 39, digital phase lock loops 40 and41, switches 42 and 43, filter design module 44, and excision filter 34.

The narrowband interference detector starts acquiring L samples beforedetection of an OFDM packet. This means that the samples provided to thenarrowband interference detector are noise and interference only with noOFDM signal present. The samples are provided to the FFT operator 36 andalso to the inputs of digital phase lock loops 40 and 41.

It should be noted that while only two digital phase lock loops areprovided in the example shown in this Figure any number M phase lockloops may be provided to suppress the M strongest interferers. Two phaselock loops are shown in this Figure so only the two strongest narrowbandinterferers will be cancelled by the interference detector of thisFigure. The N bin FFT operator 36 performs an FFT on the input samplesand performs a search on the output of the transformed data to identifythe M largest peaks from the output. The FFT operator estimates thefrequency, amplitude and phase of the M largest peaks. Once theamplitude and phase of each of the peaks are estimated as the interfereramplitude and phase, this is provided to switches 38 and 39 from theoutput of the FFT operator 36.

The phase lock loops are initialised and when the timer is set switches38 and 39 are closed to provide interferer estimates to phase lock loops41 and 40 respectively. Phase lock loops 40 and 41 each lock on to aninterferer and are continually innovated as new samples come in untileither a pilot symbol is detected or the timer times out.

If the timer times out the latest L samples are acquired by the FFToperator 36 and the process begins again.

While the phase lock loops are operating a pilot symbol may be detectedindicating the start of an OFDM packet. When a pilot symbol is detectedby pilot symbol detector 32 switches 42 and 43 are operational toprovide a lock indication and further estimates of the interfererfrequencies to filter design module 44. The filter design module thendesigns a filter for the excision filter 34 based on the estimatedinterferer frequencies from the phase lock loops and also the lockindications. If a phase lock loop provides a “no lock” on the lockindication then the estimated interferer frequency from that phase lockloop will not be taken into account in the filter design. The excisionfilter is then designed as a notch filter to remove frequenciesestimated as the interferer frequencies. Once the pilot symbol isdetected samples proceed through the pilot detector then through theexcision pilot detector 32 then through excision filter 34 to the restof the OFDM receiver 35.

FIG. 2A shows one embodiment of a narrowband interference detector ofthe invention. It is possible from the narrowband interference detectorin different ways including using a DFT instead of FFT operator 36providing a different number of phase lock loops. The particularimplementation given here should not be seen as limiting to thoseskilled in the art.

FIG. 3 shows an OFDM baseband receiver comprising four modules. Module201 includes an A/D converter driver 21, timer 22 and signal conditioner23; module 202 includes a packet detector 25, frame timer 26, narrowbandinterference suppression module 24 and first stage receiver 27; module203 includes a second stage receiver 28; and module 204 includes adecoder 23.

The data received by RF block 20 which is arranged to shift the databack to baseband. The RF block of the receiver may include a low noiseamplifier, bandpass filter, quadrature demodulator and frequencydown-converter. The baseband data is then sampled by analogue to digital(A/D) converter 21. This converts the received data from an analoguesignal to digital samples. Ideally the A/D converter samples thereceived data at greater than the nominal bit rate. The sampled signalthen passes through signal conditioner 23, that compensates for some ofthe channel and noise induced distortions. A further purpose of thesignal conditioner 23 is to digitally low pass filter the basebandsignal to remove out-of-band noise. The data may also be sampled to thenominal bit rate.

Packet detector 25 and frame timing block 26 search for the start of apacket. Packet detector 25 may also provide narrowband interferencesuppression when the packet is detected. Narrowband interferencesuppression block 24 applies narrowband interference suppression duringthe data transport phase of packet reception. In the preferredembodiment this block implements the algorithm of FIG. 2B.

Frame timing module 26 may be further arranged to provide a start ofpacket estimate to timer 22.

Once the start of a packet has been detected by packet detect block 25the packet is passed through first stage receiver 27. This receiver mayestimate and compensate for frequency and phase errors in the receiveddata. The first stage receiver also includes a Fourier transformoperator that transforms the data from time domain data to frequencydomain data. The advantage of applying the excision filter to OFDM inthe time domain before the receiver FFT is to prevent spectral leakagefrom the narrowband interference occurring at all. This greatlysimplifies the interference suppression requirements and improves theresulting BER performance.

The data is then passed into the second stage receiver 28. The secondstage receiver commences operation on frame detection. The functions ofthe second stage receiver are, initially, to estimate the time varyingsub-sample time offset and, throughout the remainder of the frame, toapply symbol timing error correction. The second stage receiver may alsobe used to update estimates of other time-varying parameters of thereceived data. The second stage receiver includes a data decision blockthat makes hard decisions on each data bit (symbol) prior to errordetection and correction.

The second stage receiver may include a demodulator. Acting togetherwith the decision process, the demodulator converts the data back from amodulation scheme, such as QPSK or 64 QAM, to binary data. After harddecisions have been made on the data the data streams are converted backfrom parallel to serial data.

Following the second stage receiver 28 is decoder 29. The decoderdecodes the coded data and performs error corrections and/or detectionup to the limit of the decoder. The decoder is matched to an encoder inthe corresponding OFDM transmitter. For example if the encoder is aReed-Solomon encoder then the decoder will be a Reed-Solomon decoder.Following decoding of the data the data is then passed to the electronicequipment attached to the receiver as data sink 30. The basic elementsof an OFDM receiver are well known and will not be discussed in moredetail.

At the completion of OFDM packet detection, the receiver returns to itsoriginal state, described by step 1 of FIG. 2B in which new interferersare searched for prior to the reception of the next in-coming OFDMpacket.

There are a number of design issues raised by the interferencesuppression technique, principally concerning the PLLs and the excisionfilter, and there is also a design trade-off between the time-outthreshold and the PLL filter bandwidth.

PLL design parameters depend on a number of system-specifics, such assub-carrier spacing and inter-packet arrival time as well as interferercharacteristics. The PLL filter bandwidth should be set to be no lessthan the OFDM sub-carrier spacing, as this is the quantisation of theinitial, DFT-based interferer carrier frequency estimate.

An example of phase lock loop operation from the simulationimplementation using a second order digital phase lock loop with loopfilter bandwidth set to 1.5 times the OFDM sub-carrier spacing is shownin FIGS. 4A to 4C. Failure of at least one PLL to achieve lock typicallyoccurred where one interferer was of significantly lower power than theother. For example when one interferer has so low power that it couldnot be distinguished from background noise. In these cases the low powerinterferer will have negligible impact on the BER. In FIGS. 4A to 4C theinitial estimates are made using a 64-point DFT. These initial valuesare shown by heavy solid lines. FIG. 4A shows the frequency estimate fora first narrowband interferer as well as the actual interfererfrequency. As can be seen the frequency estimate is close to theinterferer frequency. In this Figure the interference to noise ratio is7.5 dB. FIG. 4B shows the frequency estimate for a second narrowbandinterferer. In this Figure the interference to noise ratio is 6.6 dB. Ascan be seen the frequency estimate is close to the interferer frequency.FIG. 4C shows the operation of two phase lock loops, one for eachinterferer. After the initial estimates subsequent estimate innovationsare made using two digital phase locked loops initialised to theDFT-based estimates. A PLL ‘lock’ is said to have been achieved when thelock magnitude is less than 0.5—the lock values shown in FIG. 4Ccorrespond to the estimates shown in FIGS. 4A and 4B.

Key elements of the excision filter design include the 3 dB filterbandwidth, which should be set to being equal to the OFDM sub-carrierspacing. This is a trade-off between competing requirements, first, toprovide sufficient notch bandwidth to allow for estimation errors andtime variation in the interferer and, secondly, to restrict notchbandwidth to minimise the impact of the excision filter(s) on databearing sub-carriers which are unaffected by interference. An efficientimplementation for multiple interferes is to design a single, high-pass,excision filter, and then to (complex) frequency-shift the receivedsignal by each estimated interferer demodulated carrier frequency(negated) prior to filtering. The frequency response of a prototypefilter is shown in FIGS. 5A and 5B and was produced using two sectionsof 31 taps and 21 taps, respectively, to achieve an overall impulseresponse equivalent to that of a 165 tap conventional FIR filter.

FIG. 5A shows a one-notch excision filter frequency response and FIG. 5Bshows a two-notch excision filter frequency response. In both of theseFigures the abscissa scales are normalised to the OFDM frequency binnumber and the location of the OFDM data-bearing sub-carriers areindicated by heavy dots. In FIG. 5B the notch filter is designed tosuppress two interferers with estimated (normalised) carrier frequenciesat −2.76 and 13.5. Note that, since the overall impulse response is ofgreater duration than the guard intervals for IEEE 802.11a and IEEE802.16a, some inter-symbol interference is produced. However, since asubstantial majority of the impulse response power for a notch FIRfilter occurs around the impulse response peak, the amount ofinter-symbol interference (ISI) is small and also is deterministic. Thisprovides a further design trade-off between excision filtereffectiveness (notch depth and width) and the amount of ISI produced(which is specific to the guard interval duration of the particular OFDMsystem used.)

The efficacy of the proposed technique is established by computersimulation. The OFDM system simulated is baseband-equivalent to uncodedIEEE 802.11a, assuming perfect detection and synchronisation. Thereceiver estimated carrier frequencies for two interferers, thus twoPLLs were employed. For each simulation, two narrowband interferers wereapproximated using stationary complex cisoids having amplitudes{b₁,b₂}≐U[0,1] such that the average signal-to-interference ratio perinterferer is${SIR} = {S/\sqrt{\frac{1}{2}\left( {b_{1}^{2} + b_{2}^{2}} \right)}}$for RMS signal magnitude S, demodulated carrier frequencies{ξ₁,Ξ₂}≐U[−B/2,B/2] for OFDM passband 3 dB bandwidth B, and phases{φ₁,φ₂}≐U[−π, π]. For each transmitted packet, a signal-free trainingperiod of 1000 samples (50 μs for IEEE 802.11a) is provided to allow thePLLs to estimate the interferer carrier frequencies. FIGS. 4A to 4C showthis number of samples is sufficient to achieve a phase lock in mostcases. After 1000 samples, the excision filter is enabled in thereceiver and the OFDM packet is transmitted. Each OFDM packet comprised10,000 bits, to enable bit error rates down to 1×10⁻⁴ to be measured theresults presented have an error floor at 1×10⁻⁴.

FIG. 6A shows an example received packet magnitude, truncated to 3000samples from the start of simulation. This Figure shows the 1000 samplesignal-free training period, the effect of the excision filter on thesignal magnitude and the start of the OFDM packet. Two narrowbandinterferers are present with SIR of 10 dB and SNR of 30 dB. Theinterpolated spectrum of the first data block (first 64 samples afterthe training symbol and signal block) from the same example is shown inFIG. 6B, compared to the interference-free and unfiltered spectra forthe same data block. The effect of the excision filter on the twointerferers (at bin numbers 2.2 and 50.7, respectively) can be seen inthis example. In FIG. 6B three spectra are shown: the dashed line showsthe data symbol with noise and interference; the dotted line shows thedata symbol the noise and where the narrowband interference issuppressed using the method of the invention; and the solid line showsthe data symbol with noise and no narrowband interference.

Signal constellations for a complete OFDM packet are shown in FIGS. 7Ato 7C. FIG. 7A shows the signal constellation of the received signalplus interference and noise of FIG. 6A. FIG. 7B shows the signalconstellation of the received signal of FIG. 6A after interference isexcised using the method of the invention, and FIG. 7C shows the signalconstellation of the signal of FIG. 6A when no interference is present.The effects of interference and interference suppression on this exampleconstellation can be seen. The accuracy of the simulation BER resultscan be verified by comparison with analytical BER results, as shown inFIG. 1.

Bit error rate curves were produced by simulation, where each point on acurve is the median of 1000 simulations. This allows the effect ofvariation within the ensemble of interferer relative amplitudes, carrierfrequencies and carrier phases to be indicated by error bars on the10^(th) and 90^(th) percentiles. FIGS. 8A to 8C show mean bit error ratecurves for signal-to-interference ratios of −10 dB, 0 dB and 10 dBrespectively for the same OFDM packets with two unsuppressed narrowbandinterferes, with excised interferers, and with no interferers. Theinterferer-free curves closely follow those for conventional BPSK inAWGN.

FIG. 8A shows simulation rates of BPSK modulated OFDM with twonarrowband interferes and SIR of 10 dB. The dashed-dotted line shows thesignal plus interference and noise, the dashed line shows the filteredsignal plus interference and noise where the interference has beenexcised using the method of the invention and the solid line shows thesignal with no narrowband interference. The error bars show the 10^(th)and 90^(th) percentiles of the BER over the ensemble interferer relativepowers, carrier frequencies, carrier phases and AWGN. The same legendapplies to FIGS. 8B and 8C. It can be seen from FIGS. 8A to 8C that theeffects of narrowband interference are severe. Interference excision canbe seen to significantly improve the bit error rate for an SIR of −10dB, by reducing the median “error floor” from greater than 1×10⁻¹ toabout 1×10⁻³. However, the variation within the ensemble also can beseen to be greater for the interference excision results. This is due totwo factors: firstly, the number of data bits per OFDM block affected byexcision depends heavily on the particular values of interferer carrierfrequency and, secondly, at least one PLL failed to “lock” for at least15% of all simulations thus increasing the variability of BER results.

Close inspection of FIGS. 8A to 8C reveal that, at low SNRs, the medianexcised interferer curves are closer to the interferer-free curves atlow SIR (FIG. 8A) than at high SIR (FIG. 8C) whereas, at high SNRs, theopposite is true. This highlights that the performance-limitingmechanisms are different at low SNR than at high SNR. At low SNR,excision filter performance is limited by the accuracy of the interferercarrier frequency estimates. At high SNR, excision filter performance islimited by finite precision arithmetic effects in the filter itself.Note that the excision filters employed in these simulations introducesome ISI. It can be shown analytically, and is evident empirically, thatthe impact of this ISI on BER is negligible in comparison to the primaryperformance-limiting mechanisms discussed above for the implementationdescribed previously.

Interference excision was found to be effective across a wide range ofSIRS, as shown in FIGS. 9A to 9C. FIGS. 9A to 9C show bit error ratesfor BPSK modulated OFDM with two narrowband interferers. Each of theFigures shows three curves being the ensemble median BERs for the samedata packets where: the dashed-dotted line represents signal plus noiseand narrowband interference; the dashed line represents the signal plusnoise where the narrowband interference has been excised using themethod of the invention; and the solid line represents the signal plusnoise where no narrowband interference is present. These Figures showthat OFDM systems employing interference excision as described hereexhibit acceptable uncoded median BERs of about 1×10⁻³ for SIRs down to−30 dB. However it should be noted also that the variation within theensemble increases as the SIR decreases indicating that for someparticular values of ξT and φ, interference excision is more likely toleave a packet severely errored as the SIR decreases. FIGS. 9A to 9Calso show that the interference excision ceases to improve BERperformance when the SIR exceeds about 15 dB. This effect occurs at anSIR of about 15 dB for three reasons: first, narrowband interferencecauses increasingly fewer bit errors at higher SIRs, secondly, at agiven SIR (for lower SNRs), the low INR makes interferer carrierfrequency estimation less reliable and, thirdly, the number of biterrors introduced by the excision filter begin, as SIRs increase, toexceed the number produced by narrowband interference. For low, SIRs,these performance limitations in the interference suppression system areoutweighed by the impact of unsuppressed narrowband interference on BERperformance which otherwise would make any reliable communicationsunreliable.

A limitation of interference excision is that it provides less benefitwith QAM modulations in comparison with pure phase modulations. FIGS.10A and 10B show BER curves for 64-QAM and BPSK modulated OFDMrespectively, where the SIR and SNR have been selected so that theinterferer plus noise and interferer-free results are similar betweensystems. FIGS. 10A and 10B each include three curves. The dash-dottedcurve represents the data packet plus noise and narrowband interference,the dashed line represents signal plus noise where the interference hasbeen excised using the method of the invention, and the solid linerepresents signal plus noise where no narrowband interference ispresent. It can be seen that interference excision provides about oneorder of magnitude less benefit in BER improvement for the 64 QAM systemin comparison with the BPSK system. This is due to the excision filterattenuating (or amplifying) to some extent several OFDM data bins aroundeach interferer. This causes few bit errors in BPSK modulated OFDM asphase modulations are robust to amplitude variations. However, QAMsystems are very sensitive to amplitude variations, particularly forlarge constellations, as signal amplitude is part of the datarepresentation. Although this effect could be mitigated somewhat byequalization, use of a more robust modulation seems prudent in aninterference-limited environment in any case, as the impact ofnarrowband interference on QAM modulated OFDM also is more severe thanon BPSK modulated OFDM.

An interference suppression technique of the invention based on excisionfiltering has been described and shown by computer simulation to producea significant improvement in the bit error rate of BPSK modulated OFDMcompared to no interference suppression. Using this technique, whichworks within existing OFDM-based standards, it has been shown thatacceptable ensemble bit error rates of about 1×10⁻³ are obtainable forsignal-to-interference ratios as low as −30 dB. Excision filtering iseffective for OFDM, because it significantly reduces the majorerror-producing effect—spectral leakage—by filtering the interferencebefore the discrete Fourier transform. The interference suppressiontechnique of the invention has particular application for the datatransport phase of receiver operation rather than the detection andsynchronisation phase (although it can be used during packet detectionand synchronisation). The technique of the invention can be used incombination with a technique for narrowband interference detection andsuppression during the detection and synchronisation phases of datareception.

The foregoing describes the invention including preferred forms thereof.Alterations and modifications as will be obvious to those skilled in theart are intended to be incorporated in the scope herein as defined inthe accompanying claims.

1. A method of suppressing narrowband interference in OFDM receiversincluding the steps of; acquiring a sample of received data, estimatingparameters of each of a number of narrowband interferers from theacquired sample of data, forming an excision filter using the estimatedparameters, and inserting the excision filter into an OFDM receiver. 2.A method of suppressing narrowband interference in OFDM receivers asclaimed in claim 1 wherein the estimated parameters of the narrowbandinterferers include demodulated carrier frequency, magnitude and phase.3. A method of suppressing narrowband interference in OFDM receivers asclaimed in claim 1 wherein the step of estimating the number ofnarrowband interferers includes the steps of; performing a forward DFTon the samples, and performing a periodogram search on the output of theDFT to identify peaks in the periodogram where the number of peaks inthe periodogram corresponds to the number of interferers.
 4. A method ofsuppressing narrowband interference in OFDM receivers as claimed inclaim 1 wherein the step of estimating parameters of the narrowbandinterferers includes the steps of; estimating the frequency of aninterferer as the location of a peak on the corresponding periodogram,estimating the magnitude of the interferer as the amplitude of thecorresponding periodogram peak, and estimating the phase of theinterferer as the phase of the corresponding periodogram peak.
 5. Amethod of suppressing narrowband interference in OFDM receivers asclaimed in claim 1 including the step of initialising one digital phaselock loop for each estimated narrowband interferer using the narrowbandinterferer parameter estimates.
 6. A method of suppressing narrowbandinterference in OFDM receivers as claimed in claim 1 further includingthe step of receiving an indication of a start of packet when a datapacket is received by the OFDM receiver.
 7. A method of suppressingnarrowband interference in OFDM receivers as claimed in claim 5including the step of updating each phase lock loop each incoming sampleuntil either a counter expires or an OFDM packet is detected.
 8. Amethod of suppressing narrowband interference in OFDM receivers asclaimed in claim 7 wherein the phase locked loops are digital phaselocked loops.
 9. A method of suppressing narrowband interference in OFDMreceivers as claimed in claim 5 including the step of initialising theexcision filter with the current narrowband interferer carrier frequencyestimates from the phase locked loops that have achieved “lock’ when anOFDM packet is detected.
 10. An OFDM receiver including; a front endarranged to receive data, a data sampler arranged to provide samples ofreceived data, a narrowband interference detector that detectsnarrowband interferers in the sample of received data and estimatesparameters of each narrowband interferer, and an excision filter thatuses the estimated parameters of each narrowband interferer to reducenoise from the narrowband interferers wherein the excision filter isinserted in the OFDM receiver prior to a Fourier transform.
 11. An OFDMreceiver as claimed in claim 10 wherein the narrowband interferencedetector estimates the demodulated carrier frequency, magnitude andphase of the narrowband interferers.
 12. An OFDM receiver as claimed inclaim 10 wherein the narrowband interference detector includes a Fouriertransform operator arranged to perform a Fourier transform on thesamples and perform a periodogram search on the output of the Fouriertransform operator to identify peaks in the periodogram and at least onephase lock loop arranged to lock onto a peak identified by theperiodogram search.
 13. An OFDM receiver as claimed in claim 12 whereinthe narrowband interference detector is further arranged to estimate thefrequency of an interferer as the frequency of a peak on thecorresponding periodogram, estimate the magnitude of the interferer asthe amplitude of the corresponding periodogram peak, and estimate thephase of the interferer as the phase of the corresponding periodogrampeak.
 14. An OFDM receiver as claimed in claim 10 wherein the narrowbandinterference detector includes a timer and a filter design module. 15.An OFDM receiver as claimed in claim 10 wherein the OFDM receiver isfurther arranged to provide an estimate of the start of an OFDM datapacket to the narrowband interference detector.
 16. An OFDM receiver asclaimed in claim 15 wherein the narrowband interference detector isarranged to innovate the phase lock loop (s) until either the timertimes out or an OFDM packet is received.
 17. An OFDM receiver as claimedin claim 16 wherein the phase locked loops are arranged to estimate thecarrier frequency of the narrowband interferers.
 18. An OFDM receiver asclaimed in claim 17 wherein one phase locked loop is used for eachinterferer.
 19. An OFDM receiver as claimed in claim 10 wherein thecurrent narrowband interferer carrier frequency estimates from the phaselocked loops that have achieved “lock” are used by the filter estimatorto initialise an excision filter when an OFDM packet is detected.
 20. AnOFDM receiver as claimed in claim 10 wherein the excision filter hasimpulse response duration less than the OFDM guard interval. 21-22.(canceled)